Int. J. Communications, Network and System Sciences, 2011, 4, 468-474
doi:10.4236/ijcns.2011.47057 Published Online July 2011 (
Copyright © 2011 SciRes. IJCNS
Ultra Wi deband Microstrip Diamond Slotted Patch
Antenna with Enhanced Bandwidth
Chia Ping Lee, Chandan Kumar Chakrabarty
Centre for RF and Microwave Engineering, Department of Electronics and Communication Engineering,
Tenaga Nasional University, Kajang, Malaysia
E-mail: {ChiaPing, Chandan}
Received May 2, 2011; revised June 9, 2011; accepted June 27, 2011
An Ultra Wideband (UWB) microstrip diamond slotted patch antenna with enhanced bandwidth is presented
in this paper. The proposed antenna is simulated in CST Microwave Studio and fabricated for measurements.
Its simulated result displays impedance bandwidth from 3.28 GHz to 19.64 GHz, whereas the measured re-
sult displays the frequency region from 2.01 GHz to 18.67 GHz. The antenna complies with the return loss of
S11 < 10 dB and Voltage Standing Wave Ratio (VSWR) < 2 throughout the impedance bandwidth. Details
of the antenna design and related results such as phase angle, input impedance and radiation patterns are
discussed in this paper. This antenna has surpassed the bandwidth of UWB requirement, which is from 3.1
GHz to 10.6 GHz, and exhibits good UWB characteristics.
Keywords: Ultra Wideband (UWB), Return Loss, Impedance Bandwidth, Voltage Standing Wave Ratio
1. Introduction
Ultra Wideband (UWB) utilizes narrow pulses (on the
order of a few nanoseconds or less) for sensing and com-
munication. The Federal Communications Commission
(FCC) in the U.S.A allocated the UWB frequency spec-
trum from 3.1 GHz to 10.6 GHz below the transmitter
noise threshold of –41.3 dBm/MHz [1]. Antennas are in
high demand for various UWB applications such as wire-
less communications, medical imaging, radar and indoor
positioning [2]. This is due to its ability to enable high
data transmission rate and low power consumption. Mi-
crostrip patch antenna is frequently used in UWB an-
tenna designs due to its advantages such as lightweight,
ease of integration, small size and compact [3].
Many UWB microstrip patch antennas have been dis-
cussed in the literature to achieve the requirement for
different applications, one of which to increase the band-
width. Since microstrip patch antennas inherently have
narrow bandwidth characteristic, there have been nume-
rous techniques developed for bandwidth enhancement
in order to achieve the UWB characteristics. These an-
tennas hav e been discussed in the literature, for instance,
square-ring slot antenna [4], U-slot patch antennas [5],
dual-band slotted antenna [6], right-angle modified U-
slot antenna [7], ice cream cone antenna [8], E-shaped
patch antenna [9] and dual-band notched antenna [10].
Other techniques employed to increase the bandwidth of
antennas include meandered ground plane [11], slot
loading [12], electromagnetically coupled stacked patch
[13], patch antenna with integrated bandpass filter [14],
gap-coupled feed [15] and optimally designed impedance
matchi ng net w or k [16,17].
In this paper, the antenna is a microstrip diamond
slotted patch antenna which operates in the range of 3.28
- 19.64 GHz, thus achieving the UWB bandwidth en-
hancement. Section 2 describes the basic configuration
of the antenna design, whereas Section 3 discusses both
simulated and measured results of the antenna performa-
nces. Lastly, the findings of the simulated and measure
results are summarized in the conclusion.
2. Basic Configuration
Figure 1 illustrates the basic configuration of the an-
tenna design. The antenna is designed on an FR4 sub-
strate with the thickness of 1.6 mm and dielectric con-
stant of 3.8. The antenna consists of a larger patch with a
diamond slot, a smaller patch which serves as the feed-
line and a partial ground plane. The patch antenna’s
Copyright © 2011 SciRes. IJCNS
width and length are denoted by ‘W’ and ‘L’ respectively.
The bottom part of the patch antenna is modified into
steps denoted by ‘S1 and ‘S2 as illustrated in Figure 1.
The dimension of the diamond slot is represented by ‘sl
and ‘sw’. The feedline is denoted by ‘Fl’. The patch an-
tenna structure is printed on one side of the FR4 sub-
strate with the ground on the other side. The ground
plane is denoted by ‘Gw’ and ‘Gm’ as shown in Figure 1.
The design parameters such as the patch shape, step, the
feedline width and notched partial ground plane are op-
timized to obtain th e best return loss, S11 and impedan ce
bandwidth before determining the best dimensions for
the proposed antenna. All the simulations are carried out
using CST Microwave Studio. The dimensions of the
antenna structure are as shown in Table 1.
3. Results and Discussions
3.1. Return Loss, S11
Figure 2 illustrates the simulated and measured return
loss against frequency of the antenna. Based on the
simulated results, the antenna displays resonant frequen-
cies at 4.47 GHz with S11 of –21.86 dB, 6.34 GHz with
Figure 1. Geometry of (a) Patch antenna; (b) Ground plane.
Table 1. Dimension of antenna structure.
Basic configuration Variable Dimension (mm)
W 15.0
L 14.5
S1 1.0
S2 1.5
Patch antenna
Fl 11.5
sl 6.0
sw 9.0
Gw 2.7
Ground plane Gm 10.5
Figure 2. Simulated and measured results of return loss,
S11 (dB) against frequency (GHz).
S11 of –19.98 dB, 10.48 GHz with S11 of –29.47 dB,
12.25 GHz with S11 of –16.59 dB, 15.57 GHz with S11
of –16.07 dB and 18.13 GHz with S11 of –17.64 dB.
These frequencies are due to the patch length, L, step at
the bottom part of the patch, S2 and S1, diamond slot at
the patch antenna, notch at the ground plane and the cy-
clic reoccurrence of the first frequency, respectively. The
patch length was calculated to be around 0.67λ. As for
the frequencies due to S2 and S1, which represent the
steps at the bottom of the patch antenna, the abrupt
change in the patch antenna geometry leads to a discon-
tinuity in the microstrip line [18]. This, in fact, tunes the
capacitive coupling between the patch antenna and the
ground plane and wider impedance bandwidth is
achieved [19]. In this case, electric and magnetic field
distributions are modified near the discontinuity when
the geometry of antenna changes. The altered electric
field distribution gives rise to a change in capacitance,
and the changed magnetic field distribution can be ex-
pressed in terms of an equivalent inductance. Thus, the
discontinuity due to step S1 and S2 can be represented as
equivalent circuit as 2 stages of cascaded LC circuit as
illustrated in Figure 3. The variables can be expressed
with method of quasi-static computation as follow [18]
0.00137 1
0.258 0.8
om re
LwiH m
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Figure 3. (a) Microstrip 2 steps discontinuity; (b) Equiva-
lent circuit.
0.000987 1om re
om re
Lh nH
where Lwi for i = 1, 2 indicate inductance per unit length
of microstrip of widths W1 and W2, while Zom and εre in-
dicate the microstrip line characteristic impedance and
effective dielectric constant, respectively, and the sub-
strate thickness h is in micrometers.
Meanwhile, the resonant frequency at 15.57 GHz is
due to the notch at the ground plane. A notch at the
ground plane can be introduced to realize a series in duc-
tance. This structure has been investigated by Hoefer
[20], and it may be used to compensate for excess ca-
pacitance at discontinuities or fine-tune lengths of mi-
crostrip [18]. The configuration of the equivalent circuit
that represents the notched ground plane is as illustrated
in Figure 4. The value of the series inductance can be
obtained from the equations as follow [21]
om re
hmm z
where re
and re
are the effective dielectric con-
stants for microstrip lines with width W and (W-b), re-
spectively, and om
z and om
are the corresponding
impedances. The substrate thickness h is measured in
The simulated result displays an impedance bandwidth
with S11 below –10 dB from 3.28 GHz to 19.64 GHz.
As for the measured result, the first resonant frequency
was observed at 3.28 GHz with S11 of -20.66 dB, fol-
lowed by 5.89 GHz with S11 of –23.81 dB, 8.67 GHz
with S11 of –48.95 dB, 12.52 GHz with S11 of –25.41
dB, 15.23 GHz with S11 of –22.02 dB and 17.54 GHz
Figure 4. (a) Notch discontinuity; (b) The equivalent circuit.
with S11 of –24.08 dB. The impedance bandwidth of the
measured result covers the range from 2.01 GHz to 18.67
GHz. However, the frequency region between 4.08 GHz
to 4.7 GHz falls in S11 above –10 dB. Basically, the
measured result was slightly shifted up compared to the
simulated result, due to the loss in the SMA connector
and dielectric loss. Based on Figure 2, it is observed that
the existence of diamond slot at the patch antenna had
introduced a resonant frequency at 12.25 GHz in the
simulated result and 12.52 GHz in the measured result.
This is due to the fact that the slot cutting had taken
place in the active zone, which is the matching and ra-
diator zone. Acting on matching and radiating areas al-
lows controlling the impedance bandwid th [22]. This slot
introduces a capacitive reactance which counteracts with
the inductive reactance of the feed [23,24]. In fact, the
method of slot cutting at the patch antenna has been in-
vestigated extensively [23,25-28]. The length of the slot
was calculated to be approximately half-wavelength
(0.5λ) at its resonant frequency at 12.25 GHz. Overall,
this antenna exhibits good UWB characteristics in terms
of impedance bandwidth and return loss, with fractional
bandwidth of 142.76% in the simulated result and
120.68% in the measured result.
3.2. Phase Angle
Figure 5 illustrates the simulated and measured phase
angle against frequency of the antenna. Based on the
simulated result, it is observed that Figure 5 shows a lin-
ear response throughout the frequency region except the
range from 7.62 GHz to 13.86 GHz, in which the pulse
components in this range are radiated without distortion.
As for the measured result, the frequency region displays
linear response until 10.57 GHz. The rest of the fre-
quency region is distorted. The distortion occurred due to
Copyright © 2011 SciRes. IJCNS
Figure 5. Simulated and measured results of phase angle (˚)
against frequency (GHz).
the change in input impedance throughout the frequency
bandwidth. Overall, the phase angel pattern for this an-
tenna is satisfactory.
3.3. Voltage Standing Wave Ratio (VSWR)
Figure 6 illustrates the simulated and measured voltage
standing wave ratio (VSWR) against frequency of the
antenna. Based on the simulated result, the VSW R value
ranges from 1 to 2 throughout the frequency range. As
for the measured result, the frequency region from 4.08
GHz to 4.7 GHz and 18.68 GHz to 20 GHz displays
VSWR value above 2. Both results are validated because
the same frequency regions do fall in S11 above –10 dB
as is shown in Figure 2.
3.4. Input Impedance
Figure 7 illustrates the simulated input impedance against
frequency of the antenna. Based on Figure 7, it is ob-
served that the input impedance matching is relatively
well maintained around 50 Ohms with slight variation
throughout the frequency region. This can be validated in
Figure 6. Simulated and measured results of voltage stan-
ding wave ratio (VSWR) against frequency (GHz).
Figure 7. Simulated result of input impedance (Ohm) against
frequency (GHz).
Figure 6 as the graph in the region displays VSWR
value from 1 to 2, thus complying well with the UWB
3.5. Radiation Pattern
Figure 8 illustrates the simulated radiation patterns at
different frequencies from 3 GHz to 9 GHz at theta cut of
90˚, which is the E-plane. The radiation patterns were
simulated with the increasing frequency step of 2 GHz.
Based on Figure 8, it is observed that the radiation pat-
terns display a directional behavior, with its main lobe
direction at 0˚ and 180˚. This indicates that the concen-
tration of the field focuses on the sides of the patch.
Meanwhile, the lobes are suppressed at 90˚ and 270˚,
which originate from the front part and back part of the
patch antenna respectively. FigureS 8(b) and (c) exhibit
a similar pattern as Figure 8(a), while Figure 8(d) ex-
hibits the main lobe direction at the front patch antenna
from 0˚ to 180˚. It is observed that the directivity of the
antenna increases with increasing frequency, as the con-
centration of the field gradually focuses on the front part
of the patch antenna as is observed from Figure 8. It is
also obvious that more lobes are observed at the higher
frequency of 9 GHz. This is due to the electrically larger
size of antenna.
Meanwhile, Figure 9 illustrates the simulated radia-
tion patterns at different frequencies from 3 GHz to 9
GHz at phi cut of 90˚, which is the H-plane. The radia-
tion patterns were simulated with the increasing fre-
quency step of 2 GH z. Based on Figure 9, it is observed
that the radiation patterns exhibit an omnidirectional be-
4. Conclusions
The proposed antenna exhibits good UWB characteris-
tics, with its simulated result operating fro m 3.28 GHz to
19.64 GHz, having fractional bandwidth of 142.76%,
whereas the measured result displays frequency region
Copyright © 2011 SciRes. IJCNS
Figure 8. Simulated radiation patterns at theta cut of 90˚ at
(a) 3 GHz; (b) 5 GHz; (c) 7 GHz and (d) 9 GHz.
Figure 9. Simulated radiation patterns at phi cut of 0˚ at (a)
3 GHz; (b) 5 GHz; (c) 7 GHz and (d) 9 GHz.
Copyright © 2011 SciRes. IJCNS
between 2.01 GHz to 18.67 GHz, with fractional band-
width of 120.68%. The antenna has successfully achi-
eved enhanced UWB bandwidth, in which UWB frequ-
ency spectrum covers the range from 3.1 GHz to 10.6
GHz. Besides, it complies with the VSWR range from 1
to 2 throughout the impedance bandwidth. The phase
angle is discussed in terms of its response linearity and
distortion, whereas th e rad iation p attern s are an alyzed for
its directivity. The proposed antenna, with good UWB
characteristics and geometrically small nature, is suitab le
for wireless communication systems.
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